Typically, in electromagnetic flow meters for measuring the flow rate of a fluid that is electrically conductive, the flow rate of the fluid that flows within a measurement pipe is measured by providing a magnetic excitation electric current that alternatingly switches polarities to a magnetic excitation coil that is disposed so that the direction of the magnetic field that is produced is perpendicular to the direction of flow of the fluid that is flowing within the measurement pipe, to detect the electromotive force that is produced between a pair of electrodes that are disposed within the measurement pipe perpendicular to the magnetic field produced by the magnetic excitation coil, and sampling and performing signal processing after amplifying the electromotive force that is produced between the electrodes.
As illustrated in FIG. 5, this electromagnetic flow meter 50 is structured from a detector 50A and a converter 50B.
The detector 50A is provided with a measurement tube 51, electrodes 52, and excitation coils 53, as the primary structures thereof.
The measurement tube 51, as a whole, is made from a cylinder of a non-magnetic metal, such as stainless steel, and, on the inside thereof, has a flow path 51F for the fluid that is subject to measurement.
The excitation coil 53 are made from a pair of coils that are disposed facing the outsides of the measurement tube 51, and have a function for generating a magnetic field B, in a direction that is perpendicular to the direction of flow of the fluid that is flowing within the flow path 51F, in accordance with an excitation current Iex that is supplied from a converter 50B.
The electrodes 52 comprise a pair of electrodes that are disposed so as to contact the fluid that is flowing within the flow path 51F, on the inner wall of the measurement tube 51, facing a direction that is perpendicular to the direction of the magnetic field B that is produced by the excitation coil 53, and have a function for detecting, and outputting to the converter 50B, an electromotive force E that is produced within the fluid in accordance with the excitation of the fluid by the magnetic field B.
The converter 50B has, as its primary circuit portions, a communication interface portion 55, a signal processing portion 56, and an excitation circuit 57.
The communication interface portion 55 is connected to a higher-level device (not shown), such as a controller, through a signal line W, and has a function for generating an operating power supply from the electric power that is supplied through the signal line W from the higher-level device to supply the various functional portions, and a function for sending, to the higher-level device through the signal line W through data communication, a flow rate value for the fluid, obtained from the signal processing portion 56.
The signal processing portion 56 has a function for generating, and outputting to the excitation circuit 60, an excitation signal comprising pulse signals that have a specific excitation frequency, a function for calculating a fluid flow rate through sampling and performing signal processing based on the signal amplification and the excitation frequency, of the electromotive force E detected by the electrodes 52, and a function for outputting, to the communication interface portion 55, the flow rate value thus obtained.
The excitation circuit 60 has a function for generating, and providing to the excitation coil 53, a square wave AC excitation current for switching and controlling the excitation polarity, based on the excitation signal from the signal processing portion 56.
In this type of electromagnetic flow meter 50, a variety of noises, including electro-chemical noises, fluid noises, slurry noises, and the like, are superimposed onto the electromotive force detected by the electrodes 52. Consequently, in order to calculate the flow rate value accurately from the electromotive force, it is necessary to reduce these noises. Here these noises have the so-called “1/f” characteristic of being greater the lower the frequency band. Because of this, the higher the excitation frequency, the better the S/N ratio, and thus the more precisely the value for the flow rate can be calculated.
On the other hand, when an AC excitation current, such as comprising a square wave, in this way, is applied to the excitation coil 53, the effect of the self-inductance inherent to the excitation coil will cause the rising edge of the excitation current to be less steep, and will produce a delay in the waveform. Consequently, because the wavelength of the excitation signal is shortened when the excitation frequency is increased, the proportion of the delay in the rising edge in relation to the wavelength is increased, shortening the period of time over which an adequate magnetic field is generated, and shortening also the average steady-state region for the amplitude in the electromagnetic force that is detected by the electrode. As a result, sampling of a stable electromagnetic force becomes difficult, resulting in an increase in the error in the value of the flow rate. Because of this, it is necessary for the ramping of the excitation current to be slow, even if the excitation frequency is high.
Conventionally, there has been a proposal for a technology for improving the rising edge of the excitation current when switching the excitation polarity, through charging, into a capacitive element, the reverse induced voltage that is produced by the excitation coil, and then reusing that as electric power for excitation. See, for example, Japanese Unexamined Patent Application Publication No. H2-12221 and Japanese Patent No. 4004931).
As illustrated in FIG. 6, this type of excitation circuit 60 is structured from a switching circuit 61, a constant current circuit 62, a diode D60, a diode bridge DB, and a capacitive element C.
The constant current circuit 62 is a typical constant current circuit such as structured from an emitter-follower circuit made from, for example, a transistor Q, an op-amp OP, and a resistive element R, connected between the current output terminal Tout from the switching circuit 61 and the ground electropotential GND, where the driving current is supplied at a constant current from a power supply voltage VP to the switching circuit 61 based on a setting voltage Vcnt.
The switching circuit 61 has a function for generating, and supplying to an excitation coil L, an AC excitation current Iex through controlling polarity switching of the constant current that is supplied to the current input Tin from the power supply voltage VP by the constant current circuit 62, based on the excitation signals SA and SB, which comprise pulse signals that have mutually complementary phase relationships, outputted from a signal processing portion (not shown) of the converter.
The diode bridge DB has a function for rectifying the reverse induced voltage that is produced across (on both ends of) the terminals L1-L2 of the excitation coil L, to charge the capacitive element C. The AC terminals of the DB are connected, respectively, to L1 and L2, where the plus terminal is connected to one end of the capacitive element C, and the minus terminal is connected to the ground electropotential GND. The use of a Schottky diode structure in this DB can reduce the voltage drop in the forward direction in the individual diodes that structure the DB.
The capacitive element C is connected between the current input terminal Tin and the ground electropotential GND, and has a function for charging the reverse induced voltage that has been rectified by the DB.
The diode D60 is connected in series with the power supply voltage VP and the current input terminal Tin, and has a function for preventing back-flow, to the power supply voltage VP side, the charging voltage VC that is charged into the capacitive element C.
This switching circuit 61 is provided with four switching circuits SW 61 through SW 64 for turning the current ON/OFF, and of these, the circuit wherein SW 61 and SW 63 are connected in series and the circuit wherein SW 62 and SW 64 are connected in series are connected in parallel with each other. The terminal L1 of the excitation coil L is connected to the connection node between the contact terminal of SW 61 and the contact terminal of SW 63, and, similarly, the terminal L2 is connected to the contact node between the contact terminal of SW 62 and the contact terminal of SW 64.
As illustrated in the signal waveform diagram in FIG. 7, the excitation signals SA and SB are pulse signals, of the excitation frequency, that have a mutually complementary phase relationship, where, of these, SA controls SW 61 and SW 64, and SB controls SW 62 and SW 63.
Consequently, as shown at time T60, when there is a rising edge of SA and a falling edge of SB, SW 61 and SW 64 turn ON, and SW 62 and SW 63 turn OFF. As a result, a path is formed, as the path for the driving current that is inputted through D60 and Tin from VP, through SW 61→terminal L2→excitation coil L→terminal L1→SW 64→Tout→constant current circuit 62, to perform the polarity switching of the excitation current Iex.
On the other hand, as shown at Time T61, at the rising edge of SB and the falling edge of SA, SW 61 and SW 64 are turned OFF and SW 62 and SW 63 are turned ON. Because of this, a path is formed, as the path for the driving current, through SW 62→terminal L1→excitation coil L→terminal L2→SW 63→Tout→constant current circuit 62, to perform polarity switching of the excitation current Iex.
Here, when the polarity of the excitation current Iex is switched, the self-inductance of the excitation coil L produces a reverse induced voltage at the across-terminal voltage VL, across the ends L1-L2 of the excitation coil L. For example, at time T60, when the excitation current Iex is switched from the L1→L2 direction where it has been until this point, to the L2→L1 direction, the reverse induced voltage that is produced between the two ends L1-L2 of the excitation coil L causes the voltage at L2 to be higher than the voltage at L1. At this time, L1 is connected through SW 64 and through the constant current circuit 62 to the ground electropotential GND, and thus the high voltage generated at L2 is charged through the DB into the capacitive element C.
On the other hand, at time T61, when the excitation current Iex is switched from the L2→L1 direction where it has been until this point, to the L1→L2 direction, the reverse induced voltage that is produced between the two ends of the excitation coil L causes the voltage at L1 to be higher than the voltage at L2. At this time, L2 is connected through SW 63 and through the constant current circuit 62 to the ground electropotential GND, and thus the voltage generated at L1 is charged through the DB into the capacitive element C.
In this way, the reverse induced voltage that is generated in the excitation coil L at the time of polarity switching of the excitation current Iex is charged into the capacitive element C, and thus, during the time interval over which the charging voltage VC of the capacitive element C is higher than the voltage that is supplied from the power supply voltage VP through the diode D60, a current is supplied to the switching circuit 61 from the capacitive element C. This enables the provision of greater electric power to the Tin of the switching circuit 61, reducing the delay time from the timing of the switching of the excitation signals SA and SB until the excitation current Iex arrives at the maximum value. As a result, this makes it possible to cause the rising (or falling) of the excitation current Iex, even when at a high excitation frequency, to be faster than when compared to the case wherein the reverse induced voltage of the excitation coil L is not used (the waveform indicated by the dotted line in FIG. 7).
In this type of conventional technology, a high voltage difference is produced, through the reverse induced voltage of the excitation coil L, between the contact terminal sides of the switching circuits SW 61 through SW 64 and the control terminal side into which the excitation signals SA and SB are inputted. As explained using FIG. 6 and FIG. 7, above, a high reverse induced voltage is produced between the terminals L1 and L2 when the polarity of the excitation current Iex is switched, so the peak value of the charging voltage VC for the capacitive element C goes higher than the power supply voltage VP. For example, if the power supply voltage VP is 10 V, in some cases the charging voltage VC may increase to as much as 100 V through the reverse induced voltage within the excitation coil L.
Typically, the contact terminals to which the high voltage is applied for SW 63 and SW 64 are drain terminals of MOSFETs (N-channel), so the MOSFETs will not be damaged even if not of the high-voltage type. However, in the switches SW 61 and SW 63, typically the contact terminals to which the high voltages are applied, through the current input terminal Tin of the capacitive voltage C, are source terminals of MOSFETs (P-channel), so there is a problem in that SW 61 and SW 62 become damaged.
One strategy for avoiding damaging SW 61 through SW 64 through the high-voltage in this way is a method wherein guarding measures, such as limiting the peak value of the charging voltage VC are taken. In this method, the voltage that is charged into the capacitive element C is reduced, and thus there is a problem in that the reverse induced voltage of the excitation coil L is not used effectively as electric power for driving the excitation coil.
Moreover, as another method by which to prevent damage to SW 61 and SW 62 through the high-voltage in this way, one may consider the use of a high-voltage switching circuit made from a MOSFET that has adequate voltage durability performance, that is, from a high-voltage MOSFET, so as to not cause damage even if the reverse induced voltage of the excitation coil L is applied to these switches SW 61 and SW 62. As an example, while the absolute maximum rating (VDS) for the drain-source voltage in a normal MOSFET is about 20 V, in a high-voltage MOSFET, the VDS is about 100 V, or even greater than 100 V.
However, high-voltage MOSFETs tend to have higher resistance, where the excitation current Iex is produced by this resistance, and, as a result, there is a problem in that the excitation coil cannot be driven efficiently by the power supply voltage VP.
Moreover, as described above, the voltage on the contact terminal side, which is that which is turned ON and OFF by the MOSFET that structures the switching current, varies greatly, from the 10 V that is the power supply voltage VP to 100 V, which is the peak value of the charging voltage VC.
On the other hand, when a high-voltage is turned ON or OFF in a high-voltage MOSFET, a high-voltage must be applied to the gate terminal in combination with this voltage. Because of this, in order to control a voltage that has a broad scope of variability, the voltage at the gate terminal side must also be switched, and this has a problem in that it makes the control system extremely complex.
The present invention is to solve such problems, and an aspect thereof is to provide an excitation coil for an electromagnetic flow meter wherein the excitation current can ramp up quickly, through effectively using the reverse induced voltage of the excitation coil L, while avoiding damaging the switching circuit through high-voltage.